Method and receiver apparatus for determining a correlation value considering frequency correction

ABSTRACT

A receiver apparatus for determining a correlation value of a predetermined repetitive chip sequence in a receive signal, the chip sequence having a chip sequence duration and a chip sequence duration cycle. The receiver apparatus has a receive unit for receiving the receive signal and a segmenter for providing receive signal segments from the receive signal, two receive signals representing the same chip sequence in different repetition cycles and having at least one repetition cycle time spacing.

BACKGROUND OF THE INVENTION

The present invention is in the field of frequency correction of receivesignals, as e.g. occurring as Doppler shifts in spread spectrum receivedsignals when there is a motion between a transmitter and a receiver.

In wireless communications it is a known problem that frequency shiftsoccur, if receiver and transmitter are not stationary. The so-calledDoppler shifts are a general problem of mobile communications, asreceivers of frequency shifted signals often need to apply frequencycorrections in order to achieve sufficient signal quality. This problemhas a general character and occurs in basically most of mobilecommunication systems. In the following, reference will be made tospread spectrum communication systems, however, similar considerationscan be taken with respect to other mobile communication systems as well.

In spread spectrum communication systems, predefined sequences,so-called chip sequences, are utilized in order to spread an informationsignal in the frequency domain. At a receiver, a replica of said chipsequence is generated, in order to correlate the generated sequence withthe receive or received signal. In the following the expressions ofreceive and received signal will be used synonymously. Throughcorrelation, the desired signal can be extracted from the receivesignal, this operation is also referred to as despreading. The utilizedchip sequences are also known as PRN codes (PRN=Pseudo Random Noise).The correlation at the receiver can wipe out the PRN code and isbasically done by correlation, i.e. by multiplying and integrating theincoming signal with the code replica.

In order to enable proper correlation, the beginnings of both codes,i.e. the replica and the code within the received signal, should bealigned in time, which can be done by an iterative search throughcorrelations with received signals of different time shifts. So in afirst dimension a proper time shift can be evaluated.

Another dimension of search can be frequency, where signal errors ordistortions are due to the so-called Doppler effects. Due to the mutualmotion of the transmitter and receiver, the received signal can beshifted in frequency. Therefore, the second search dimension may be infrequency. Both effects, frequency and time shifts, are unknown at thereceiver, and different algorithms can be used to decrease them.

Finding the delay of the incoming code, also referred to as the codedelay, is the aim of for example GNSS (GNSS=Global Navigation SatelliteSystems). A correlation in the frequency domain, for example utilizingthe FFT (FFT=Fast Fourier Transform), can be used to determine thisvalue. As mentioned above, care also needs be taken on the Dopplereffect, as the frequency shift should be deleted or decreased as well.In order to do so, one known algorithm is to perform an iterative searchthrough a grid of possible Doppler frequencies, which is determined bythe Doppler frequency range, which can for example be [−5, 5] kHz in aGNSS with static receivers.

Moreover, there are known algorithms for increasing asignal-to-noise-ratio (SNR) of a correlation value at the receiver. Inthe following correlation of receive signal segments with chip sequenceswill be illuminated.

Here and for the remainder a correlation value shall correspond to aresult of correlating two sequences, which yields a correlation functionor sequence composed of multiple correlation values. However, withinthese resulting sequences there may be particular correlation values,e.g. a peak value or a value fulfilling a predetermined condition whichis to be detected for example to find a correct delay of a receivedsequence. Therefore the expression of correlation value may correspondto a value of a correlation result in terms of a sequence or function.

An actual public algorithm for increasing the SNR is for example theso-called alternate half-bit method (AHBM) and the so-called full-bitmethod (FBM). They defer with respect to the coherent integration time,which is used at a correlator for integration, and which is half of abit (AHBM) or a full-bit (FBM) duration respectively. Integration timeswhich exceed those durations, refer to incoherent integrations, becauseoutside these durations the modulating data is unknown, thereforecoherent superposition of these signal parts may not be anticipated.However, it is also known that the gain of incoherent integration is notas high as the gain of coherent integration, especially in low SNRscenarios. For example in indoor scenarios, where rather weak receivesignals can be received, with a short coherent integration time properdetection may not be possible. A problem of these conventional systemsis that a proper positioning within indoor scenarios or low SNRscenarios is difficult.

Furthermore, the amount of coherently integrated signal duration limitsthe grid side of a Doppler frequency grid to be searched. The maximumfrequency jump, which can be done, is given by the expression

$\begin{matrix}{{\Delta \; f_{bin}} = {\frac{2}{3 \cdot T_{COH}}.}} & (1)\end{matrix}$

This condition takes into account, that phase variations within acoherently integrated signal fraction (of duration T_(COH)) should belimited in order to prevent destructive superposition. If one demandsthat at least half of the signal power can be integrated within thecoherent time, it yields that the maximum allowable phase shift istwo-thirds of π, implying the above condition for the maximum frequencyjump.

For example, in indoor scenarios, it is already known that the SNR isvery low. Therefore, it seems to be appropriate to increase the coherentintegration time in order to acquire s satellite's signal properly andfor afterwards obtaining the users position more properly as well.However, according to the above condition, the higher the coherentintegration time the finer the Doppler frequency grid, and consequently,the higher the complexity since a finer Doppler grid provides a lot moreDoppler frequencies to be considered. Moreover, when extending coherentintegration times to a higher value than a single bit duration, thisimplies that the data symbols or bits within the coherent integrationtime should be known to be able to combine them coherently. Otherwise ifthe bits are unknown and integrating across bit boundaries of differentbits, bit sign changes can cause destructive superposition. This againwould imply that a pilot channel has to be available in thecommunication system, i.e. a known transmit data sequence. If thetransmitted bit sequence is not known, and noting that one bittranslates at least one chip sequence or a number of chip sequenceswhich could be combined coherently, the receiver does not know whether asign change has occurred or not at the bit boundaries. Therefore,coherent combining across bit borders is very critical if the bits areunknown.

As already mentioned above, increasing the coherent integration timegenerates a proportional increment of the number of correlations to beperformed, assuming that at least one correlation needs to be performedper Doppler frequency shift conceivable. Moreover, even if a system hasa pilot channel the pilot channel consisting of a repeating datasequence would then imply that a synchronization process has to becarried out, which can dramatically enlarge the complexity of adetection algorithm in terms of the amount of operations that need to becarried out.

The abovementioned AHBM and FBM algorithms have only a limited coherentintegration time, due to the bit boundaries and therefore theirperformance is very limited with respect to low SNR scenarios. Forlonger coherent integration times a pilot channel is necessitated, buteven if there is a pilot channel, the number of operations that have tobe performed with the pilot channel may take a lot of processing time atthe receiver. If such synchronization is necessitated a receiverstructure may become more complex, its power consumption may rise andthe complexity of the detection algorithm may increase significantly.

WO 006/119816a1 describes a concept for decoding a signal based on anincoming stream of data samples representing at least one downconverteddigitized spread spectrum source signal. The received data samples aresubdivided into a number of data blocks, which are individuallycorrelated with a locally available code replica, before being processedfor Doppler frequency compensation.

US 2007/0025476 A1 discloses methods and apparatus for determiningcarrier frequency errors of a serial offset quadrature pulse shapedsignal, such as a minimum shift keyed signal. The carrier frequencyerror is determined by receiving a quadrature pulse shaped signal havinga synchronization sequence detecting synchronization of the quadraturepulse shaped signal and storing a baseband inphase signal and a basebandquadrature signal of the synchronization sequence while detectingsynchronization. After detecting synchronization segments of the storedbaseband inphase and quadrature signals are read and correlated with thespreading sequence. Carrier frequency error is then estimated based onphase differences between each of the correlated signals.

U.S. Pat. No. 6,195,328 B1 provides an improved acquisition and trackingsystem for GPS signals. The system relies on block adjustment of thesynchronizing signal of the bi-phase shift keying signal in order toobtain correct carrier frequency and phase angle. This improved systemhas the advantages of being more robust in the presence of noise thanconventional approaches and also of lending itself to simplifiedimplementation since synchronization of the coarse/acquisition code needonly be within half of a chip in order to maintain lock.

SUMMARY

According to an embodiment, a receiver apparatus for determining acorrelation value of a predetermined repetitive chip sequence in areceive signal, the chip sequence having a chip sequence duration and achip sequence repetition cycle may have a receive unit for receiving thereceive signal; a segmenter for providing receive signal segments fromthe receive signal, two receive signal segments representing the samechip sequence in different repetition cycles and having at least onerepetition cycle time spacing; a frequency corrector for determiningsets of frequency corrected receive signal segments based on sets ofcorrection frequencies, the correction frequencies being based on thechip sequence duration and the chip sequence repetition cycle; a chipsequence generator for generating the predetermined chip sequence; acorrelator for correlating the predetermined chip sequence with thefrequency corrected receive signal segments of a set to acquire a setcorrelation value; and a selector for selecting one of the setcorrelation values meeting a predetermined condition as the correlationvalue.

According to another embodiment, a method for determining a correlationvalue of a predetermined repetitive chip sequence in a receive signal,the chip sequence having a chip sequence duration and a chip sequencerepetition cycle may have the steps of receiving the receive signal;providing receive signal segments from the receive signal, two receivesignals segments representing the same chip sequence in differentrepetition cycles and having at least one repetition cycle time spacing;determining sets of frequency corrected receive signal segments based onsets of correction frequencies, the correction frequencies being basedon the chip sequence duration and the chip sequence repetition cycle;generating the predetermined chip sequence; correlating thepredetermined chip sequence with the frequency corrected receive signalsegments of a set to acquire a set correlation value; and selecting oneof the set correlation values meeting a predetermined condition as thecorrelation value.

According to another embodiment, a computer program may have a programcode for performing, when the program code runs on a computer, a methodfor determining a correlation value of a predetermined repetitive chipsequence in a receive signal, the chip sequence having a chip sequenceduration and a chip sequence repetition cycle having the steps ofreceiving the receive signal; providing receive signal segments from thereceive signal, two receive signals segments representing the same chipsequence in different repetition cycles and having at least onerepetition cycle time spacing; determining sets of frequency correctedreceive signal segments based on sets of correction frequencies, thecorrection frequencies being based on the chip sequence duration and thechip sequence repetition cycle; generating the predetermined chipsequence; correlating the predetermined chip sequence with the frequencycorrected receive signal segments of a set to acquire a set correlationvalue; and selecting one of the set correlation values meeting apredetermined condition as the correlation value.

The present invention is based on the finding that a more efficientcorrelation value or function can be retrieved, respectively a higherSNR with a similar computational burden, if shorter repetitive sequencesof the receive signals are used for coherent combining, where the timespacing between the repetitive sequences is larger than the sequenceduration itself. In embodiments these repetitive sequences maycorrespond to a pilot channel having repetitive symbols, where a symbolmay be represented by a chip sequence such as for example a pseudorandom noise sequence (PRN). This way, the number of frequency offsetsto be taken into account in order to compensate for the Doppler shiftscan be lowered, reducing the computational complexity. Moreover, thenumber of correlations to be performed within an iterative search amongall possible Doppler offsets or frequencies can be further reduced, if arepetition cycle of the sequence, i.e. the time spacing between therepetitive segments as e.g. a repetition period of a pilot symbol, andthe sequence duration are chosen in a manner that groups of possibleDoppler frequencies occur, wherein within a group of possible Dopplerfrequencies correlations can be approximated by phase shifts of thecorrelation value of, for example, a center frequency of the group.Looking at the scheme in another way, a better SNR can be achieved atthe same computational complexity than with conventional approaches.Moreover, correlation values can be further improved if Dopplerfrequency changes are taken into account as well. Especially if therepetition cycle, respectively the number of sequences taken intoaccount being separated by a repetition cycle, is long, the Dopplerfrequency, i.e. frequency offsets, may change during the acquisition ofthe repetitive sequences. SNRs of correlation values or functions can befurther improved when taking the Doppler frequency changes into account.

In some embodiments, e.g. for GNSS, the Doppler frequency change ratecan be taken into account. Because this effect can produce changes onthe phases e.g. of the correlation peaks of each segment. Thisconsideration may produce a third-search-dimension, i.e. a frequencychange rate search. In embodiments a frequency change rate may also beapproximated by phase shifts and the computational complexity may befurther reduced.

BRIEF DESCRIPTION OF THE DRAWINGS

In the following, embodiments of the present invention will be describedusing the accompanying figures, in which

FIG. 1 shows a block diagram of an embodiment of a receiver apparatus;

FIG. 2 illustrates a received signal comprising a repetitive chipsequence;

FIG. 3 illustrates a coarse and a fine grid of correction frequencies;

FIG. 4 illustrates consideration of frequency changes;

FIG. 5 shows a flowchart of an embodiment of a method for determining acorrelation value;

FIG. 6 illustrates the relation between correlations and phase shifts;and

FIG. 7 illustrates an embodiment of a parallel receiver structure.

DETAILED DESCRIPTION OF THE INVENTION

FIG. 1 shows a block diagram of a receiver apparatus 100. The receiverapparatus comprises a receive unit 110, a segmenter 120, and frequencycorrector 130, a chip sequence generator 140, a correlator 150, and aselector 160.

Here and for the remainder a correlation value shall correspond to aresult of correlating two sequences, which yields a correlation functionor sequence composed of multiple correlation values. However, withinthese resulting functions or sequences there may be particularcorrelation values, e.g. a peak value or a value fulfilling apredetermined condition, which is to be detected for example to find acorrect delay of a received sequence. Therefore the expression ofcorrelation value may correspond to a value of a correlation result interms of a sequence or function.

The receiver apparatus 100 is adapted for determining a correlationvalue or function of a predetermined repetitive chip sequence in areceive signal 115, which is illustrated between the receive unit 110and a segmenter 120 in FIG. 1. The receive signal 115 is shown along atime axis, on which chip sequences 1, 2, 3, . . . , N are pointed out.The chip sequences have a chip sequence duration and a chip sequencerepetition cycle, e.g. like a pilot data sequence. In other words, fromthe receive signal 115 in FIG. 1, it can be seen that a chip sequencehas a duration and that a chip sequence repeats after a certainrepetition cycle, e.g. in a repetitive pilot symbol frame, wherein apilot symbol is represented by, respectively modulating, one ormultiple, possibly also repetitive, PRN sequences, i.e. the segments inthe receive signal 115 which are also labeled with 1, 2, 3, . . . , Nare the same repetitive chip sequences.

The receiver apparatus 100 comprises the receive unit 110 for receivingthe receive signal 115. The segmenter 120 is adapted for providingreceive signal segments from the receive signal, two receive signalsegments representing the same chip sequence in different repetitioncycles and having at least one repetition cycle time spacing. Inembodiments the receive signal segments can correspond to PRN sequences,which are modulated by repetitive pilot symbols. These segments are thenprovided to the frequency corrector 130, which is adapted fordetermining sets of frequency corrected receive signal segments based onsets of correction frequencies, the correction frequencies being basedon the chip sequence duration and the chip sequence repetition cycle.

The chip sequence generator 140 is adapted for generating thepredetermined chip sequence within the receiver apparatus 100. Thepredetermined chip sequence corresponds to a local replica of therepetitive chip sequence in the receive signal. The correlator 150 isadapted for correlating the predetermined chip sequence with thefrequency corrected receive signal segments of a set to obtain a setcorrelation value or function; and the selector 160 is adapted forselecting one of the set correlation values or functions meeting apredetermined condition as the correlation value or function.

In order to better illustrate the components of the receive signal 115,FIG. 2 shows another embodiment of a receive signal comprising arepetitive chip sequence.

FIG. 2 shows two time lines, one at the bottom and one at the top, onwhich a receive signal is indicated. The receive signal's magnitude isindicated by |r|. From the bottom diagram it can be seen that a receivesignal has repetitive segments, which have a repetition cycle or timespacing T_(SPACE)=KT_(PILOT), where K is a positive integer value andwherein repetitive segments are indicated by repetitive patterns. At thetop of FIG. 2, a magnified version of a segment is shown, where theduration of one segment is T_(sub-piece). From the top diagram of FIG.2, it can be seen that one segment is composed of a chip sequence, whichin embodiments may be composed of several chip sequences as well.

Referring back to FIG. 1, it is the task of the segmenter 120 to providereceive signal segments from the receive signal, wherein two receivesignal segments represent the same sequence in a different repetitioncycle, having at least one repetition cycle time spacing. In embodimentsthe receive signal segments can correspond to PRN sequences, which aremodulated by repetitive pilot symbols. Moreover in embodiments more thantwo receive signal segments may be considered, which all represent thesame sequence in a different repetition cycle or pilot period, having atleast one repetition cycle or period duration time spacing. In otherwords, at the receiver apparatus 100, an attempt to find the time stampsof a certain sequence within the receive signal is made by correlatingthe receive signal segments, wherein the said chip sequence is present,and by adding up said correlations in order to improve an SNR, e.g. inorder to distinguish a peak value within said correlations.

FIG. 1 shows that the segmenter 120 provides the receive signal segmentsr₁, r₂, . . . , r_(N) to the frequency corrector 130. It is pointed outthat it is an example to use N segments. Embodiments are not restrictedto any number of segments.

The frequency corrector 130 now corrects the frequency of the receivesignal segments, by applying different sets of correction frequencies tothe set of segments. In other words, a set of correction frequenciescomprises one correction frequency per receive signal segment. Said onecorrection frequency is then applied to the corresponding segment of thereceive signal segments, yielding a set of frequency corrected receivesignal segments. The frequency corrected receive signal segments of theset of frequency corrected receive signal segments are then individuallycorrelated with the predetermined chip sequence and the results aresummed up, yielding a set correlation value. It is to be noted thatcorrelation value corresponds to one value of a correlation sequence orfunction, i.e. correlation sequences or function may be summed up.

The set of correction frequencies comprises correction frequencies,which are based on the chip sequence duration and the chip sequencerepetition cycle. In other embodiments, the correction frequencies mightalso have to take into account the receive signal's frequency changes.In the embodiment described in FIG. 1, each frequency correction setcomprises one correction frequency for each receive signal segment. Thefrequency corrector 130 can be adapted for determining one set ofcorrection frequencies per frequency of a coarse grid of frequenciesbetween a minimum coarse frequency and a maximum coarse frequency, witha coarse step size depending on the chip sequence duration. The numberof different sets of correction frequencies then corresponds to thenumber of different correction frequencies of the coarse grid, eachfrequency correction set comprises one correction frequency for eachreceive signal segment.

FIG. 3 illustrates a coarse and a fine grid of correction frequencies.At the top of FIG. 3, a frequency line with a coarse grid of frequenciesf_(Ci) is illustrated, where i is an index for the coarse correctionfrequencies, which can be of any value in embodiments. The coarse gridof frequencies may start at the minimum coarse frequency f_(C1) 310 andreach to a maximum coarse frequency 320. Within the grid, the coarsefrequencies may be equidistant, e.g. as indicated by the coarsefrequency step size 330, which can depend of the chip sequence duration.In one embodiment, the minimum coarse frequency 310 would be greaterthan or equal to −5 kHz, the maximum coarse frequency 320 could be lessthan or equal to 5 kHz, and the coarse step size 330 could be less thanor equal to two-thirds divided by the chip sequence duration, in orderto ensure coherency between the correlation values of each frequencycorrected segment of one set.

Moreover, the frequency corrector 130 may be adapted for determining oneset of correction frequencies f_(Ci,j) per frequency of a fine grid ofcorrection frequencies around a correction frequency f_(Ci) of thecoarse grid, where j is an index for the fine correction frequencies,which can be of any value in embodiments. This is indicated in the lowerpart of FIG. 3, where an area around a coarse correction frequencyf_(C3) is magnified. The diagram at the bottom on FIG. 3 shows afrequency axis, with a fine grid of correction frequencies f_(C3,−1),f_(C3,0), f_(C3,1), exemplified around one correction frequency f_(C3)of the coarse grid. Similar to what was said above, the fine grid cancomprise a minimum fine frequency offset 350 and a maximum finefrequency offset 360, where offset refers to a respective centerfrequency, which may be a frequency from the coarse frequency grid.Moreover, a fine step size 370 may be applied for equidistant fine gridcorrection frequencies.

In one embodiment, the minimum fine frequency offset 350 may be greaterthan or equal to −0.5 divided by the chip sequence repetition cycle, themaximum fine frequency offset 360 may be less than or equal to 0.5divided by the chip sequence repetition cycle and the fine step size 370may be less than or equal to one-third divided by the chip sequencerepetition cycle.

In other embodiments, the frequency corrector 130 may be adapted fordetermining one set of correction frequencies per frequency change rateof a frequency change rate grid, the frequency change rate grid having aminimum frequency change rate, the maximum frequency change rate and afrequency change rate step size. Similar considerations as weredescribed above would result for the frequency change rates in suchembodiments.

FIG. 4 illustrates the considerations of frequency change rates. FIG. 4shows a time line, on which several receive signal segments areindicated by labels 1, 2, . . . , N. If transmitter and receiver moverelatively to each other, then a frequency offset occurs. In FIG. 4 itis assumed that this frequency offset f₁ applies to segment 1. Now, ifthe relative velocity between the transceiver and the receiver changesin time, then said frequency offset changes as well. Assuming that therelative velocity between the transmitter and the receiver constantlyincreases, i.e., the acceleration between them is constant, forequidistant receive signal segments, the same frequency offsets yield.In FIG. 4, it is indicated that between receive signal segment 1 and 2,a frequency change of Δf occurs, and thus,

f ₂ =f ₁ +Δf.

Accordingly, for receive signal segment N yields

f _(N) =f ₁+(N−1)Δf.

In this embodiment it was assumed that an acceleration betweentransmitter and receiver is constant. Embodiments are not limited tothis assumption, which serves for explanatory purposes only. Generally,any arbitrary accelerations, motions or velocities between a transmitterand a receiver may be taken into account, in order to derive a set ofcorrection frequencies. The correction frequencies may then changeindependently between the segments along the time line.

In one embodiment, the minimum frequency change rate may be greater thanor equal to −2 Hz/sec, the maximum frequency change rate may be lessthan or equal to 2 Hz/sec and the change rate step size for thefrequency change rate grid between the minimum and maximum frequencychange rates may be less than or equal to 0.5 Hz/sec.

Thus, in embodiments the number of sets of correction frequencies may bedetermined the number of coarse correction frequencies in the coarsegrid multiplied by the number of fine correction frequencies in the finegrid multiplied by the number of different frequency change rates in thefrequency change rate grid.

In embodiments, the correlator 150 may be adapted for correlating thepredetermined chip sequence with each frequency corrected receive signalsegment of a set to obtain a segment correlation value or function foreach frequency corrected signal segment of the set of correctionfrequencies. The correlator 150 can then be further adapted forcombining all segment correlation values or functions of the set toobtain one set correlation value per set of correction frequencies. Thenumber of different sets of correction frequencies can be determined bythe number of coarse correction frequencies multiplied by the number offine correction frequencies multiplied by the number of differentfrequency changes that can be taken into account.

In embodiments the frequency corrector 130 may be adapted for providingphase shifts per correction frequency of the fine grid of correctionfrequencies to frequency correct correlation values of frequencycorrected receive signal segments based on the coarse grid offrequencies. Referring back to FIG. 3, instead of determiningcorrelations for each set of correction frequencies for each correctionfrequency of the fine grid, the fine grid correlations may be replacedby coarse grid correlations multiplied by phase shifts. The frequencycorrector 130 may then be adapted for providing phase shifts also perfrequency change rate of the frequency change rate grid to frequencycorrected receive signal segment correlations based on the coarse gridof frequencies. In other words, in embodiments the correlations for thefine grid frequencies and the frequency rate changes may be replaced orapproximated by applying phase shifts to correlations of frequencycorrected receive signal segments having been corrected according tocorrection frequencies of the coarse grid.

The application of the phase shifts, e.g. the multiplication of thecorrelation values or functions resulting from the coarse grid, may becarried out in the correlator 150. In other embodiments it may becarried out by a separate approximator, which receives sets of segmentcorrelation values from the correlator 150 and applies sets of phaseshifts to the sets of segment correlation values. A set correlationvalue or function can then be determined by the approximator, or thecorrelator 150 respectively, by combining the phase shifted segmentcorrelation values of a set. In other words, the frequency corrector 130may provide sets of phase shifts instead of the correction frequenciesof the fine grid and the frequency change grid. Based on combining thephase shifted segment correlation values or functions of a set offrequency corrected receive signal segments according to the coarsefrequency correction grid, the set correlation values of the fine andthe frequency change rate frequency correction grids can be determined.In embodiments these operations may be carried out by the correlator150, the frequency corrector 130 or an approximator.

The received signal 115 may be a Code-Spread-Spectrum receive signal ora CDMA receive signal (CDMA=Code Division Multiple Access). In otherembodiments, it may be a receive signal according to a GNSS, GPS(GPS=Global Positioning System), the Galileo system, GLONASS(GLONASS=Russian GNSS, etc.). However, embodiments are not limited tothese systems, their scope may refer to any frequency distorted orshifted receive signal.

FIG. 5 illustrates a flow chart of an embodiment of a method fordetermining a correlation value of a predetermined repetitive chipsequence and a receive signal, the chip sequence having a chip sequenceduration and a chip sequence repetition cycle. The first step 510 isreceiving the receive signal. Step 510 is followed by the step 520 ofproviding receive signal segments from the receive signal, two receivesignal segments representing the same chip sequence in differentrepetition cycles and having at least one repetition cycle time spacing.A step 530 follows, wherein the sets of frequency corrected signalsegments based on sets of correction frequencies are determined, thecorrection frequencies being based on the chip sequence duration and thechip sequence repetition cycle. In a step 540, the predetermined chipsequence is generated. In the following step 550, correlating thepredetermined chip sequence with the frequency corrected receive signalsegments of the set to obtain a set correlation value is carried out. Ina last step 560, one of the set correlation values meeting apredetermined condition is selected as the correlation value.

In embodiments the predetermined condition, which is met by the selectedcorrelation value could be the highest correlation value from acorrelation function. In other embodiments in order to determine acorrelation value with a certain certainty, the correlation value mayhave to exceed a predetermined threshold. This could, for example, be inorder to determine a correct correlation value in a corresponding timewith a given certainty, i.e. for example a bit error ratio, e.g. 2%.Related to the bit error ratio there could be a certain SNR thresholdand as soon as a correlation value above the threshold is determined,the procedure is ended. Naturally, a number of different criteria inorder to select the correlation value are conceivable, the inventivemethod shall not be limited to any particular one of them.

In general, the aim of carrying out or evaluating the Doppler grid, i.e.the coarse, fine and change rates, is to wipe out the Doppler Effectduring the coherently integrated signal piece. The remaining Dopplerfrequency, which can be called f_(D) ^(error), is the wave that is stillmodulating the signal. In order to enable coherent superposition, when acontinuous signal piece is used, this frequency should be small enoughso to approximately have the same phase in all coherently addedcorrelation peaks. As explained above, this is what leads to thecondition of the Doppler bin size of

$\begin{matrix}{{\Delta \; f_{bin}} = \frac{2}{3 \cdot T_{COH}}} & (2)\end{matrix}$

Since embodiments of the present invention segment the coherentintegration time, this condition can be avoided and therefore a higherfrequency bin size realized. This advantage results from embodiments notintegrating a continuous T_(COH) signal piece, but several signalpieces, corresponding to sub-pieces, similar to a time hopping manner.In other words, embodiments may utilize a signal of T_(COH) length,formed by equally time spaced repetitive sub-pieces. The length of thetime pieces can be a multiple of the PRN code duration, i.e. a chipsequence duration.

If the receive signal provides a pilot data sequence, i.e. a knownsequence, the time space T_(SPACE) between the sub-pieces can be amultiple of the duration of this sequence, T_(PILOT) in order to ensurethat the data symbols on the sub-pieces are the same. This isillustrated in FIG. 2, where the repetition cycle of the sequence isT_(SPACE) and the chip sequence duration is T_(sub-piece). Embodimentstherefore achieve two advantages, first a coherent integration sum canbe carried out due to the equality of symbols of the sub-pieces andsecond, no pilot sequence synchronization is needed, sinceT_(SPACE)=K·T_(PILOT), wherein K is a natural number greater than orequal to 1. Thus, receive signal segments may be taken into account,which have a spacing that is an integer multiplier of the repetitioncycle.

In order to perform a coherent sum between sub-pieces correlation peaks,the phase of the remaining Doppler frequency f_(D) ^(error) on thosepieces should be almost “equal”. Two conditions can be formulated, ifthe sub-pieces or segments are separated as described above, the Dopplerbin definition changes through the following conditions:

$\begin{matrix}{{\Delta \; f_{bin}^{coarse}} = \left. \frac{2}{3 \cdot T_{{sub} - {piece}}}\Leftrightarrow{{\; f_{D}^{error}} < \frac{\Delta \; f_{bin}^{coarse}}{2}} \right.} & (3) \\{{{\; f_{D}^{error}} = \frac{K}{T_{SPACE}}},} & (4)\end{matrix}$

where K is a whole number.

With the first condition, it can be ensured that a correlation peak iscorrectly generated within the coarse Doppler bin, which is nearest tothe real Doppler frequency f_(D), in every sub-piece or segment,although it may not be possible to distinguish it due to the low SNR. Inother words the first condition refers to coherent superposition of thesignals within one receive signal segment, yielding the coarsecorrection frequency grid.

With the second condition, it can be ensured that the phases of the notdistinguishable correlation peaks of each sub-piece or segment areequal, and therefore, they can be coherently added. In other words, thesecond condition refers to coherent superposition of the resultingcorrelation values, sequences or functions of different receive signalsegments, yielding the fine frequency correction grid. To reach thistarget, another Doppler grid has to be done on the surroundings of theprevious correlated Doppler bin, i.e. a fine frequency correction gridis applied on top of the coarse frequency correction grid. This fineDoppler grid range can be defined by T_(SPACE) in the following way:

$\begin{matrix}{{{Fine\_ Doppler}{\_ Range}} = {f_{D}^{{coarse}\_ {bin}} + {\left\lbrack {\frac{- 1}{2 \cdot T_{SPACE}},\frac{1}{2 \cdot T_{SPACE}}} \right\rbrack.}}} & (5)\end{matrix}$

As can easily be seen, the size of the range can be decreased whenT_(SPACE) is increased. Embodiments may use this fact as being guided toa computational burden reduction. Furthermore, if this fine Dopplerrange is small enough, no correlations have to be necessarily performedwith respect to the fine frequency correction grid, as thosecorrelations can be approximated by a phase change applied on the coarsefrequency correction correlations.

FIG. 6 illustrates this idea by showing a frequency axis with coarsefrequencies indicated by stars grid with a coarse frequency step size ofΔf_(bin) ^(coarse). Around each of the coarse correction frequencies,there are a number of fine correction frequencies which are indicated bybars of the fine Doppler range. Instead of evaluating correlations ofeach of the frequencies of the fine Doppler range, correlations may beperformed only for the coarse correction frequencies (stars), the finecorrection frequencies can be approximated by phase shifts, as isindicated on the right hand side of FIG. 6.

In embodiments, the approximation may be performed as follows:

correlation(f _(D) ^(fine) ^(—) _(bin))=correlation(f _(D) ^(coarse)^(—) _(bin))·exp(Δφ)  (6)

where Δφ=−j2π(f_(D) ^(fine) ^(—) _(bin)−f_(D) ^(coarse) ^(—)_(bin))·n·T_(SPACE), n is the signal sub-piece index, f_(D) ^(fine) ^(—)_(bin) is the bin inside of the fine Doppler range and f_(D) ^(coarse)^(—) _(bin) is the coarse Doppler bin within the current fine Dopplerrange.

As mentioned above, the higher T_(SPACE) is chosen the longer it takesfor constructing a complete set of sub-pieces or segments of sizeT_(COH). In some scenarios this could be critical, if the Dopplerfrequency changes during this time. Embodiments may therefore also takethe frequency change rate into account, which can be denoted by:

$\begin{matrix}{\frac{\partial f_{D}}{\partial t}.} & (7)\end{matrix}$

This parameter, for a normal static GNSS receiver may for examplenecessitate a value within the range of [−1, 1] Hz/sec. This value mayrepresent the changes in velocity between the receiver and thetransmitter, i.e. an acceleration value. The longer the considered timeperiod, the higher the impact of the change rate of the frequency. Thechange in frequency relates directly to the remaining Doppler frequency,as

$\begin{matrix}{\left( {\frac{\partial f_{D}^{error}}{\partial t} = \frac{\partial f_{D}}{\partial t}} \right),} & (8)\end{matrix}$

i.e. f_(D) ^(error) may not be a constant value, but a linearly varyingvalue:

$\begin{matrix}{f_{D}^{error} = \left. f_{D}^{error} \middle| {}_{0}{{+ \frac{\partial f_{D}}{\partial t}} \cdot {t.}} \right.} & (9)\end{matrix}$

In this embodiment f_(D) ^(error) may be approximated as a linearlyvarying value, according to the above Tailor sequence. Generally, inembodiments a more complex variation of the frequency error can be takeninto account.

In order to combat the transmitter-receiver acceleration effect, anothergrid, the frequency change grid, can be taken into account along therange of

$\begin{matrix}{\frac{\partial f_{D}}{\partial t}.} & (10)\end{matrix}$

In embodiments this grid may represent a third dimension search andinvolves an increment of computational burden. In other embodiments, itmay also be carried out within the same approximation as alreadymentioned above. Actually, a small change can be introduced on the phaseapproximation expression in order to take this third grid into account:

$\begin{matrix}{{\Delta \; \phi} = {{{- j}\; 2{{\pi \left( {f_{D}^{{fine}\_ {bin}} - f_{D}^{{coarse}\_ {bin}}} \right)} \cdot n \cdot T_{SPACE}}} - {j\frac{\partial f_{D}^{bin}}{\partial t}{\pi \cdot \left( {n \cdot T_{SPACE}} \right)^{2}}}}} & (11)\end{matrix}$

The computational complexity of this method again depends on the gridstep size, which is now determined by the last two dimensions, i.e. thefine Doppler frequency grid, f_(D) ^(fine) ^(—) _(bin) and the Dopplerfrequency change rate or grid

$\begin{matrix}{\left( \frac{\partial f_{D}^{bin}}{\partial t} \right).} & (12)\end{matrix}$

In embodiments, once the sub-pieces size, i.e. the chip sequenceduration and T_(SPACE) are fixed, these steps only depend on T_(COH),i.e. on the number of sub-pieces that are desired to integratecoherently. The higher this number, the finer the grids can be done.

In other embodiments, a reduction of the whole operation can be achievedif more than one complex multiplier are implemented in hardware in orderto carry out more multiplications simultaneously. As shown above in theΔφ definition, an approximation may be applied to every sub-piece orsegment, wherein the approximation value may depend on the sub-pieceindex n, Δφ[n]. A parallelization may reduce the computational time bythe number of multipliers implemented in parallel. FIG. 7 illustrates anembodiment of a receiver structure having such a parallelization. FIG. 7shows n_(MAX) frequency corrected receive signal segments 702, 704 and706, as e.g. provided by a receive unit 110, a segmenter 120 and afrequency corrector 130. It is supposed in FIG. 7 that all of thesereceive signal segments have already been corrected coarse according toa coarse correction frequency f_(D) ^(coarse) ^(—) _(bin), i.e. in thisembodiment the steps described in the following may be carried out foreach correction frequency of the coarse correction frequency grid, i.e.for each of the frequency corrected sets of receive signal segments foreach correction frequency of the coarse frequency grid.

Each of the receive signal segments 702, 704 and 706 is then correlatedin a correlator 150 with the locally generated predetermined chipsequence, yielding the segment correlation values or functions 712, 714and 716. Instead of correlating also with the fine correctionfrequencies, the outputs of the segment correlation values or functions712, 714, and 716 are multiplied with phase shifts, within themultipliers 722, 724 and 726. The phase shifts Δφ[1], Δφ[2], . . . ,Δφ[n_(MAX)] correspond to the above definition, i.e. there may be asmany phase shifts as the product of number of different frequency changerates and number of different frequency bins within the fine frequencycorrection grid. FIG. 7 shows an embodiment wherein the application ofthe phase shifts is done in the correlator 150. Other embodiments maycarry these operations out in a separate approximator. Correspondingphase shifts may be provided by the frequency corrector, respectively byanother separate entity, e.g. the approximator.

The phase shifted segment correlation values are then added in the adder730, upon which a correlation value respectively a correlation functionyields, which can be provided to a detector, respectively, selector inorder to find the correlation value, e.g. a correlation peak. From theresult displayed in FIG. 7 at the output

In order to better appreciate the improvements of the embodiments, acomparison of the computational complexity between the conventional wayand an embodiment is considered in the following, based on the GalileoGNSS system. This system provides a pilot channel and a pilot datasequence of 25 BPSK (BPSK=Binary Phase Shift Keying) symbols. The unitfor weighting the computational complexity will be the number of complexmultiplications to be performed by each method. First, the mainparameters of the Galileo system that affect the acquisition algorithmsare shown. Some of them also depend on the receiver's parameters andthey have been fixed nowadays to the most commonly chosen values,however embodiments are not restricted to them.

The coherently integrated time desired T_(COH)=100 ms. The GalileoParameters are PRN length of 4 ms, the number of points in 4 ms is

N=2¹⁴(16K FFTs)

and

T_(PILOT)=100ms,

made of a sequence of 25 BPSK symbols.

The classic algorithm uses a Radix-2 FFT module in hardware.

The parameters of the new algorithm are

$\begin{matrix}{T_{SPACE} = \left. {100\mspace{14mu} {ms}}\rightarrow{{Fine\_ Doppler}{\_ Range}} \right.} \\{{= {f_{D}^{coarse\_ bin} + {\left\lbrack {5,5} \right\rbrack \mspace{14mu} {Hz}}}},}\end{matrix}$ $\begin{matrix}{T_{{sub} - {piece}} = {{one}\mspace{14mu} {PRN}\mspace{14mu} {code}}} \\{{= {4\mspace{14mu} {ms}}},}\end{matrix}$ Δ f_(D)^(fine) = 0.08  Hz${\Delta \left( \frac{\partial f_{D}}{\partial t} \right)} = {0.3\mspace{14mu} {Hz}\text{/}{\sec.}}$

These grids have been calculated with a small script, based on the Δφexpression given above and provide a minimum efficiency of 84.5% on thecoherent integration, due to the phase mismatch of the signalsub-pieces. Therefore, in order to obtain a gain of 100 ms coherentintegration, 116 ms will have to be integrated (T_(COH) ^(efficient)),in order to ensure this gain. This way, the number of sub-pieces to beintegrated raises to N_(SUB-PIECE)=29, and the number of multipliers(N_(multipliers)) implemented will also be 29.

The parameters for the classic algorithm are

${{\Delta \; f_{D}^{bin}} = {\frac{2}{{3 \cdot 100}\mspace{14mu} {ms}} \approx {6\mspace{20mu} {Hz}}}},\begin{matrix}{{Number\_ Correlations} = {\frac{T_{COH}}{PRN\_ length} \cdot \frac{10\mspace{14mu} {KHz}}{\Delta \; f_{D}^{bin}}}} \\{= {\frac{100\mspace{14mu} {ms}}{4\mspace{14mu} {ms}} \cdot \frac{10\mspace{14mu} {KHz}}{6\mspace{14mu} {Hz}}}} \\{\approx 41667.}\end{matrix}$

Each correlation is done with the 3 FFTs and the number ofmultiplications in one radix-2 FFT is given by

${\frac{N}{2} \cdot {\log_{2}(N)}},$

where N is the number of points of the FFT. In this case, N=2¹⁴.Moreover, a disadvantage of this method is that it necessitatessynchronization with the pilot symbol sequence, which in this case, ismade of 25 symbols. Therefore, all these operations need to bemultiplied by 25.

Hence

${{Computational\_ Burden}\mspace{14mu} ({CB})} = {25_{syn} \cdot 3_{{FFT}/{corr}} \cdot {Number\_ Correlations} \cdot \frac{N}{2} \cdot {\log_{2}(N)}}$

CB=358.4·10⁹ complex multiplications.

For the new algorithm (3-dimension search), follows

$\mspace{20mu} {{{\Delta \; f_{D}^{coarse}} = {\frac{2}{{3 \cdot T_{{sub} - {piece}}}\mspace{11mu}} \approx {166\mspace{20mu} {Hz}}}},\mspace{20mu} \begin{matrix}{{Number\_ Correlations} = {\frac{T_{COH}^{efficient}}{PRN\_ length} \cdot \left\lceil \frac{10\mspace{14mu} {KHz}}{\Delta \; f_{D}^{coarse}} \right\rceil}} \\{= {\frac{116\mspace{14mu} {ms}}{4\mspace{14mu} {ms}} \cdot \frac{10\mspace{14mu} {KHz}}{166\mspace{14mu} {Hz}}}} \\{{\approx 1747},}\end{matrix}}$${{{Number\_ Approximation}{\_ per}{\_ Coarse}{\_ Doppler}{\_ Bin}} = {N_{app}=={\frac{{Fine\_ Doppler}{\_ Range}}{\Delta \; f_{p}^{fine}} \cdot \frac{{Doppler\_ Change}{\_ Rate}{\_ Range}}{\Delta \left( \frac{\partial f_{D}}{\partial} \right)}}}},\mspace{20mu} \begin{matrix}{N_{app} = {\left\lceil \frac{{5\mspace{14mu} {Hz}} - \left( {{- 5}\mspace{14mu} {Hz}} \right)}{0.08\mspace{14mu} {Hz}} \right\rceil \cdot \left\lceil \frac{{1\mspace{14mu} {Hz}\text{/}\sec} - \left( {1\mspace{14mu} {Hz}\text{/}\sec} \right)}{0.3\mspace{14mu} {Hz}\text{/}\sec} \right\rceil}} \\{{= 875},}\end{matrix}$${{CB} = {{{3_{{FFT}/{corr}} \cdot {Number\_ Correlation} \cdot \frac{N}{2}}{\log_{2}(N)}} + {\left\lceil \frac{10\mspace{14mu} {KHz}}{\Delta \; f_{D}^{coarse}} \right\rceil \cdot \frac{N_{{sub}\_ {piece}} \cdot N_{app} \cdot N}{N_{multipliers}}}}},\mspace{20mu} {{CB} \approx {{1.46 \cdot 10^{9}}{complex}\mspace{14mu} {{multiplications}.}}}$

The gain on the computational burden with the new algorithm is easilyappreciated, because it has been reduced by a factor of 245.

Embodiments of the present invention provide the advantage that theyincrease a SNR in order to acquire satellites on GNSS. Embodiments maybe used in all CDMA or other communication systems, in which an unknownDoppler effect takes place. This may be the case for example in CDMAbased GNSS, like GPS, Galileo, Compass, etc. Embodiments may also beused on spread spectrum communication systems like for example Glonass(Russian GNSS), which even if it is not a CDMA system, it does usePRN-code for spreading its signal spectrum.

Embodiments of the present invention enable the breaking of the law ofDoppler frequency bins, in terms of the number of correlations to beperformed. The coherently integrated signal times may be extended out ofthe bit boundaries, even with a lower computational complexity thanbefore. Embodiments provide another significant advantage, in that theymay not rely on a synchronization with a pilot sequence, as therepetition period or cycle of pilot sequences is known in such systems.Embodiments may therefore enable the acquisition in indoor scenarios ofGNSS stand-alone receivers, which can be easily equipped with anembodiment and yielding an acceptable time consumption.

Depending on certain implementation requirements of the inventivemethods, the inventive methods can be implemented in hardware or insoftware. The implementation can be performed using a digital storagemedium, in particular a disc, DVD, CD, etc., having electronicallyreadable control signals stored thereon, which cooperate with aprogrammable computer system such that the inventive methods areperformed. Generally, the present invention is therefore, a computerprogram product with a program code stored on a machine-readablecarrier, the program code being operative for performing the inventivemethods when the computer program product runs on a computer. In otherwords, the inventive methods are, therefore, a computer program having aprogram code for performing at least one of the inventive methods whenthe computer program runs on a computer.

While this invention has been described in terms of several embodiments,there are alterations, permutations, and equivalents which fall withinthe scope of this invention. It should also be noted that there are manyalternative ways of implementing the methods and compositions of thepresent invention. It is therefore intended that the following appendedclaims be interpreted as including all such alterations, permutationsand equivalents as fall within the true spirit and scope of the presentinvention.

1-16. (canceled)
 17. Receiver apparatus for determining a correlationvalue of a predetermined repetitive chip sequence in a receive signal,the chip sequence comprising a chip sequence duration and a chipsequence repetition cycle, comprising a receive unit for receiving thereceive signal; a segmenter for providing receive signal segments fromthe receive signal, two receive signal segments representing the samechip sequence in different repetition cycles and comprising at least onerepetition cycle time spacing; a frequency corrector for determiningsets of frequency corrected receive signal segments based on sets ofcorrection frequencies, the correction frequencies being based on thechip sequence duration and the chip sequence repetition cycle; a chipsequence generator for generating the predetermined chip sequence; acorrelator for correlating the predetermined chip sequence with thefrequency corrected receive signal segments of a set to acquire a setcorrelation value; and a selector for selecting one of the setcorrelation values meeting a predetermined condition as the correlationvalue.
 18. Receiver apparatus of claim 17, wherein the frequencycorrector is adapted for determining the correction frequencies based onreceive signal frequency changes.
 19. Receiver apparatus of claim 17,wherein each frequency correction set comprises a correction frequencyfor each receive signal segment.
 20. Receiver apparatus of claim 17,wherein the frequency corrector is adapted for determining one set ofcorrection frequencies per frequency of a coarse grid of frequenciesbetween a minimum coarse frequency and a maximum coarse frequency with acoarse step size being based on the chip sequence duration.
 21. Receiverapparatus of claim 20, wherein the minimum coarse frequency is greaterthan or equal to −5 kHz, the maximum coarse frequency is less than orequal to 5 kHz and the coarse step size is less than or equal totwo-thirds divided by the chip sequence duration.
 22. Receiver apparatusof claim 20, wherein the frequency corrector is adapted for determiningone set of correction frequencies per frequency of a fine grid offrequencies around a correction frequency of the coarse grid, the finegrid comprising a minimum fine frequency offset from the coarsecorrection frequency, a maximum fine frequency offset from the coarsecorrection frequency and a fine step size.
 23. Receiver apparatus ofclaim 22, wherein the minimum fine frequency is greater than or equal to−0.5 divided by the chip sequence repetition cycle, the maximum finefrequency offset is less than or equal to 0.5 divided by the chipsequence repetition cycle and the fine step size is less than or equalto one-third divided by the chip sequence repetition cycle.
 24. Receiverapparatus of claim 17, wherein the frequency corrector is adapted fordetermining one set of correction frequencies per frequency change rateof a frequency change rate grid, the frequency change rate gridcomprising a minimum frequency change rate, a maximum frequency changerate and a frequency change rate step size.
 25. Receiver apparatus ofclaim 24, wherein the minimum frequency change rate is greater than orequal to −2 Hz/s, the maximum frequency change rate is less than orequal to 2 Hz/s and the change rate step size is less than or equal to0.5 Hz/s.
 26. Receiver apparatus of claim 17, wherein the correlator isadapted for correlating the predetermined chip sequence with eachfrequency corrected receive signal segment of a set to acquire a segmentcorrelation value for each frequency corrected receive signal segmentfor a set of correction frequencies and for combining all segmentcorrelation values of each set of correction frequencies to acquire onecorrelation value per set of correction frequencies.
 27. Receiverapparatus of claim 22, wherein the frequency corrector is adapted forproviding a phase shift per correction frequency of the fine grid forthe frequency corrected receive signal segments based on the coarse gridof frequencies.
 28. Receiver apparatus of claim 24, wherein thefrequency corrector is adapted for providing a phase shift per frequencychange rate of the frequency change rate grid for the frequencycorrected received signal segments based on the coarse grid offrequencies.
 29. Receiver apparatus of claim 27, wherein the correlatoris adapted for applying the phase shift to the segment correlationvalues and for combining the phase shifted segment correlation values toacquire one correlation value per set of correction frequencies. 30.Receiver apparatus of claim 17, wherein the receive unit is adapted forreceiving a CDMA—(CDMA=Code Division Multiple Access), a GNSS(GNSS=Global Navigation Satellite System), a GPS (GPS=Global PositioningSystem), a Galileo, a Glonass-signal.
 31. Method for determining acorrelation value of a predetermined repetitive chip sequence in areceive signal, the chip sequence comprising a chip sequence durationand a chip sequence repetition cycle comprising: receiving the receivesignal; providing receive signal segments from the receive signal, tworeceive signals segments representing the same chip sequence indifferent repetition cycles and comprising at least one repetition cycletime spacing; determining sets of frequency corrected receive signalsegments based on sets of correction frequencies, the correctionfrequencies being based on the chip sequence duration and the chipsequence repetition cycle; generating the predetermined chip sequence;correlating the predetermined chip sequence with the frequency correctedreceive signal segments of a set to acquire a set correlation value; andselecting one of the set correlation values meeting a predeterminedcondition as the correlation value.
 32. A computer readable mediumstoring a computer program comprising a program code for performing,when the program code runs on a computer, a method for determining acorrelation value of a predetermined repetitive chip sequence in areceive signal, the chip sequence comprising a chip sequence durationand a chip sequence repetition cycle comprising: receiving the receivesignal; providing receive signal segments from the receive signal, tworeceive signals segments representing the same chip sequence indifferent repetition cycles and comprising at least one repetition cycletime spacing; determining sets of frequency corrected receive signalsegments based on sets of correction frequencies, the correctionfrequencies being based on the chip sequence duration and the chipsequence repetition cycle; generating the predetermined chip sequence;correlating the predetermined chip sequence with the frequency correctedreceive signal segments of a set to acquire a set correlation value; andselecting one of the set correlation values meeting a predeterminedcondition as the correlation value.